1. Field of the Invention
The present invention relates to high-efficiency power supplies and similar devices.
2. Description of the Related Art
A multi-phase, parallel resonant converter is a good choice for high-efficiency, high-power DC/DC applications, such as telecommunication power supplies and similar applications. Load current sharing is a key issue in such applications. Interleaved, parallel power converters can provide an output with a small ripple. However, interleaved, parallel power supplies need additional metal-oxide-semiconductor field-effect transistors (MOSFETs), and therefore, the cost of interleaved, parallel power supplies is higher, and an additional gate-drive circuit is needed. The dynamic performance of interleaved, parallel power is not very good when the load is changing. In particular, at light loads, interleaved, parallel power can be inefficient because of switching losses of all of the MOSFETs.
Known LLC resonant converters are attractive for isolated DC/DC applications, such as flat-panel TVs, laptop adapters, server computers, etc. because of their attractive features: smooth waveforms, high efficiency, and high power density. Known LLC resonant converters have been widely used due to the high efficiency as a result of zero-voltage switching (ZVS) of the primary-side MOSFETs and of zero-current switching (ZCS) of the secondary-side diodes in which the secondary-side diodes are switched between current-flowing and current-blocking states so that the diode current decreases to zero before the next half period. For high-power applications, the current stress on the power devices increases with the power rating. Connecting multiple converters, or phases or stages, in parallel is a good technique to address this problem of current stress. But, because of the tolerances of resonant components, the resonant frequency of each individual converter will be different. Thus, the output currents of the different phases will be different. A small component tolerance, e.g., such as less than 5%, can cause significant current imbalance as shown, for example, in FIG. 4. Therefore, current sharing is needed to achieve multiphase operation.
FIG. 30 shows a known two-phase converter with phases 1 and 2. Each phase includes a transformer with primary and secondary windings. The transformer turns ratio is n. A primary circuit is connected to the primary winding, and a secondary circuit is connected to the secondary winding.
The primary circuit of phase 1 includes primary switches Q11, Q12 connected in series and includes resonant inductor Lr, resonant capacitor Cr, and magnetizing inductor Lm connected in series. The magnetizing inductor Lm is connected in parallel with the primary winding. The current iLr1 is the resonant current in phase 1. The primary circuit of phase 2 includes primary switches Q21, Q22 connected in series and includes resonant inductor aLr, resonant capacitor bCr, and magnetizing inductor cLm connected in series. The values a, b, c indicate that the resonant parameters for these two phases are different. The magnetizing inductor cLm is connected in parallel with the primary winding. The current iLr2 is the resonant current in phase 2. The primary circuits of phases 1 and 2 are connected to the voltage input Vin.
The secondary circuit of phase 1 includes a rectifying stage including synchronous rectifiers SR11, SR12 connected to the secondary winding and an output capacitor Co1 connected to the rectifying stage. The current irect1 is the current through the rectifying stage. The current io1 is the load current of phase 1. The secondary circuit of phase 2 includes a rectifying stage including synchronous rectifiers SR21, SR22 connected to the secondary winding and an output capacitor Co2 connected to the rectifying stage. The current trect2 is current through the rectifying stage. The current io2 is the load current of phase 2. The secondary circuits of phases 1 and 2 are connected to the output Vo. The current io is the output current. Resistance Ro represents the resistance of the load.
A mathematic model of the LLC converter is needed for analyzing the current sharing characteristics. For simplicity, a two-phase LLC converter without using a sharing method is shown in FIG. 30. FIG. 31 is the equivalent circuit based on fundamental harmonic analysis (FHA). In steady-state, the load resistor Ro is separated Ro1 and Ro2 according to each load current io1, io2. The primary-side equivalent ac resistors Rac1, Rac2 are:
                    {                                                                                                  R                                          o                      ⁢                                                                                          ⁢                      1                                                        =                                                            1                      k                                        ⁢                                          R                      o                                                                      ,                                                      R                                          o                      ⁢                                                                                          ⁢                      2                                                        =                                                            1                                              (                                                  1                          -                          k                                                )                                                              ⁢                                          R                      o                                                                      ,                                  k                  ∈                                      [                                          0                      ,                      1                                        ]                                                                                                                                                                R                                          a                      ⁢                                                                                          ⁢                      c                                                        =                                                                                    8                        ⁢                                                  n                          2                                                                                            π                        2                                                              ⁢                                          R                      o                                                                      ,                                                      R                                          a                      ⁢                                                                                          ⁢                      c                      ⁢                                                                                          ⁢                      1                                                        =                                                                                    8                        ⁢                                                  n                          2                                                                                            π                        2                                                              ⁢                                          R                                              o                        ⁢                                                                                                  ⁢                        1                                                                                            ,                                                      R                                          a                      ⁢                                                                                          ⁢                      c                      ⁢                                                                                          ⁢                      2                                                        =                                                                                    8                        ⁢                                                  n                          2                                                                                            π                        2                                                              ⁢                                          R                                              o                        ⁢                                                                                                  ⁢                        2                                                                                                                                                    (        1        )            where k is the impedance sharing error that is between 0 and 1. If k=0.5, then the load power is equally shared by the two phases. If k=0 or 1, then the load power can only be provided by one of the phases.
Three known current-sharing methods have been used with multiphase LLC converters. The first known current-sharing method is the active method which adjusts the equivalent resonant capacitor or inductor to compensate for the components' tolerances using additional MOSFETs as shown in FIGS. 27 and 28. This method can achieve excellent load-sharing performance. An example of this known method using a switched capacitor is shown in FIG. 27.
The known current-sharing method using switched capacitors shown in FIG. 27. Each phase has a switched capacitor. The switched capacitor includes the capacitor Cs with two transistors connected in series with each other and connected in parallel across the capacitor Cs. The two transistors define an additional switch that charges or discharges the capacitor Cs. The equivalent capacitor is a variable capacitor with a changing duty ratio.
The known current-sharing method using a variable inductor is shown in FIG. 28. The converter in FIG. 28 is similar to the converter in FIG. 27, except that the switched capacitor is replaced with variable inductors Lst1, Lst2. The variable inductors Lst1, Lst2 include an extra circuit with additional switches that control the coupled windings of the variable inductors Lst1, Lst2.
This known current-sharing method uses an additional circuit, which includes switches, a passive element such as a capacitor or an inductor, and a detecting current circuit. The circulated current can be controlled by changing the resonant frequency based on the additional circuit. The equivalent resonant inductance or capacitance is changed by the variable inductor or the switched capacitor in the additional circuit. Thus, the resonant frequency is changed as the inductance or capacitance is changed. These known current-sharing methods with the switched capacitor and the variable inductor suffer from high cost, complex control, and inferior dynamic performance because of the required sensing circuit and of the need to control the additional switches.
A second known current-sharing method is the DC-voltage, self-balanced method that uses series DC-bus capacitors as shown in FIG. 29. The series DC-bus capacitors of the two-phase converter shown in FIG. 29 includes two capacitors C1, C2 connected in series, which can share the current by automatically adjusting the voltage of the two series capacitors C1, C2. Capacitor C1 is connected in parallel across primary switches Q11, Q12, and capacitor C2 is connected in parallel across primary switches Q21, Q22. The two large series DC capacitors C1, C2 are connected in series to share the input DC voltage. FIG. 29 shows a two-phase LLC converter to explain the principle. The mid-point voltage is changed according to the power of the two phases. The input voltage of the first module is the voltage of the capacitor C1, and the input voltage of the second module is the voltage of the capacitor C2. The input voltage of each module can be changed to balance power by the series DC capacitor. The output voltage is same for each of the modules; thus, the current can be shared. Thus, the converter has low cost and good load-current sharing performance.
To balance the capacitor voltage, it is better to use a two-phase LLC converter. It is difficult to use additional modules. It is hard to achieve a modular design with the second known current-sharing method because the DC voltage stress is reduced as the number of modules increases. The total input voltage and output voltage is constant. When two modules are used in the series DC capacitor current-sharing method, the input voltage of each of the modules is about half of the total input voltage. When three modules are used, the input voltage of each module is about a third of the total input voltage. When the input voltage is low, the design of the LLC converter will not be optimized because the resonant current (i.e., the input current) will be increased. In addition, when one module fails, the input voltage for the other modules will have a large change, which is not desirable.
A third known current-sharing method is based on a three-phase, three-wire structure for three-phase LLC converters based on a 120°-phase-shift method, which has good load-current sharing near the resonant frequency as all of the three-phase resonant currents are zero. But this third known current-sharing method is only suitable for three LLC converter phases connected in parallel. The load current will not share with more than three phases.
Therefore, the known current-sharing methods do not provide cost effective, flexible current sharing for multi-phase LLC resonant converters.